Method for measuring a gap between a proximity probe and a conductive target material

ABSTRACT

A digital eddy current proximity system including a digital impedance measuring device for digitally measuring the proximity probes impedance correlative to displacement motion and position of a metallic target object being monitored. The system further including a cable-length calibration method, an automatic material identification and calibration method, a material insensitive method, an inductive ratio method and advanced sensing characteristics.

This application is a divisional patent application of U.S. Ser. No.10/042,514, filed Jan. 8, 2002, now U.S. Pat. No. 6,664,782, which is adivisional patent application of U.S. Ser. No. 09/425,830, filed Oct.22, 1999, issued Feb. 12, 2002 as U.S. Pat. No. 6,346,807.

FIELD OF THE INVENTION

The instant invention relates generally to a digital impedancemeasurement systems and, in particular, to a digital eddy currentproximity system for analyzing and monitoring rotating and reciprocatingmachinery.

BACKGROUND OF THE INVENTION

Analog eddy current proximity systems which analyze and monitor rotatingand reciprocating machinery are known in the art. These analog systemstypically include a proximity probe located proximate a target object(e.g., a rotating shaft of a machine or an outer race of a rollingelement bearing) being monitored, an extension cable and analogconditioning circuitry. The target, proximity probe (a noncontactingdevice which measures displacement motion and position of an observedconductive target material relative to the probe), extension cable andconditioning circuitry components are all designed to interact in such away that a voltage output from the circuitry is directly proportional toa distance between the probe and the target. This distance is commonlyreferred to as “gap”.

The interaction that takes place between these components is in accordwith the following rules: First, the electrical impedance measured atthe conditioning circuitry is the electrical combination of the target,the probe including an integral sensing coil and cable, the extensioncable and the conditioning circuitry. This impedance is usually calledthe “Tank Impedance” or parallel impedance (Zp). Second, this tankimpedance is linearized and converted into a voltage directlyproportional to gap. Third, the conditioning circuitry measuresimpedance at a specific frequency that is a function of its owncircuitry. Generally, the circuitry runs at the frequency where thereactive component of the tank impedance approaches zero. In otherwords, the circuitry is a resonant system, so the frequency of operationwill be where the phase shift of the impedance is approximately zerodegrees. In reality, the phase shift is not exactly zero due to, interalia, manufacturing and component variations and tolerances of eachanalog system.

In order to compensate for these variations and tolerances, each analogsystem is required to be calibrated to have a parallel impedance whichis as close as possible to a predefined ideal parallel impedance whileremaining substantially unsusceptible to the multitude of variations andtolerances found in the target, probe, extension cable, and conditioningcircuitry. Simultaneously, each analog system is calibrated to have amaximum sensitivity to changes in gap. Moreover, each system isgenerally required to be calibrated to monitor one specific targetmaterial.

These analog systems are also generally burdened by temperaturevariations in the target, the probe including the integral sensing coiland cable, the extension cable and the conditioning circuitry due to thesevere temperature variations in rotating and reciprocating machineryenvironments. Thus, each system is required to be designed around amultitude of component tolerances to compensate for the severetemperature variations engendered in these environments. Furthermore,these analog systems must also be designed around the sensitivity tochanges in the conductivity and permeability of the target, the sensingcoil, and the cable, which can greatly effect the precision of thesesystems.

Moreover, interchangeability problems arise from variations in thetarget, probe, extension cable, and conditioning circuitry which causethe tank impedance (Zp) versus gap to vary slightly from nominalresulting in a proclivity towards, inter alia, variations in incrementalscale factor (ISF), variations in average scale factor (ASF) anddeviations from a straight line (DSL). The incremental scale factor(ISF), variations in average scale factor (ASF) and deviations from astraight line (DSL) are common ways to specify transducer performance asis well known in the art.

It is critical that the displacement motion or position between thetarget and the sensing coil of the proximity probe remains within thelinear range of the proximity probe for providing accurate and reliablemeasurements over a wide range of circuit and environmental conditionsin order to operate rotating and reciprocating machinery safely andefficiently. Heretofore, the ability to provide accurate and reliablemeasurements over a wide range of circuit and environmental conditionshas been dependent on, inter alia, designing and manufacturing eachproduction unit within close tolerances and going through laboriouscalibration methods to compensate for the circuit and environmentalconditions.

For the foregoing reasons, there is a need for an eddy currenttransducer system that, inter alia, substantially eliminates themanufacturing and component variations and tolerances of the prior artanalog systems, a system that provides correct gap reading for differenttarget materials and a system which is easy to calibrate.

Additionally, there is a need to solve the general problem ofcompensating for temperature errors, temperature profiles of differenttarget materials and changes in component conductivity and permeabilityin order to preclude anomalous behavior in eddy current transducersystems.

Furthermore, there is a need for an eddy current transducer system thathas better linearity and interchangeability. Moreover, there is a needfor an eddy current transducer system that does not require componentchanges when re-calibrated to a new or different target material.

SUMMARY OF THE INVENTION

The instant invention is distinguished over the known prior art in amultiplicity of ways. For one thing, the instant invention provides aunique digital system for digitally measuring an unknown electricalimpedance. Additionally, the instant invention provides a digitalproximity system that is a direct one for one replacement for existinganalog eddy current proximity systems which is compatible with anyexisting (or future) eddy current proximity probe (a noncontactingdevice which measures displacement motion and position of an observedconductive or metallic target material relative to the probe) andextension cable assembly. Thus, the instant invention can directlyreplace the analog conditioning circuitry of prior art analog systemsthereby eliminating the anomalies associated with manufacturing andcomponent variations, and tolerances of these systems. Furthermore, theinstant invention eliminates the laborious design and calibrationmethods required to calibrate prior art analog systems in order tocompensate for manufacturing and component variations, tolerances andenvironmental conditions.

In one form, the instant invention provides a system which includes aunique voltage ratio apparatus and method for digitally measuring anunknown electrical component value. The system accomplishes this bydigitizing a first voltage impressed across a serial coupling of a firstelectrical component and a second electrical component, and bydigitizing a second voltage impressed across the second electricalcomponent only. Each of the two digitized voltages is then convolvedwith digitized waveforms to obtain a first and a second complex voltagenumber. A ratio of the second complex number to a difference between thefirst and the second complex number is determined and multiplied by aknown value of the first electrical component to determine the unknownvalue of the second electrical component. A resistance means having aknown value can be employed as the first electrical component. Thesecond electrical component can take the form of a proximity probehaving an unknown impedance value which, when determined by the instantinvention, can be correlated to a distance between the probe and ametallic target object being monitored by the probe. Iterativelyrepeating the voltage ratio method results in continuously digitallydetermining the unknown impedance values of the probe which can bedirectly correlated to the continuous displacement motion and positionof the target being monitored relative to the probe. In one form, thedigitally determined impedance values can be transformed into analogsignals and used to trip alarms, circuit breakers, etc., when thesignals are outside nominal operating ranges set by plant operators.

Additionally, the instant invention provides a system that can be usedas a direct one for one replacement for existing (and future) analogeddy current proximity systems. The system includes a unique apparatusand method for digitally measuring the impedance of a proximity probeand an extension cable (if employed) which includes the unique voltageratio apparatus and method delineated supra to obtain an unknownimpedance of the proximity probe. Then, the system mirrors a circuitequivalent impedance of an existing (or future) analog proximity circuitand combines the measured impedance with the circuit equivalentimpedance for defining a parallel or tank impedance. The defined tankimpedance is then correlated to a distance between the probe and ametallic target object being monitored by the probe. Hence, the systemcan continuously digitally determine the unknown impedance value of theprobe by iteratively repeating the aforementioned method and thencorrelate the digitally measured probe impedance values to thecontinuous displacement motion and position of the target for analyzingand monitoring rotating and reciprocating machinery.

More particularly, the instant invention provides a system which employsat least one eddy current proximity probe having a multi-axial probecable coupled to a sensing coil located proximate a conductive target tobe monitored. The sensing coil is coupled to ground and to a secondterminal of a resistor via the probe cable and an extension cable (ifemployed). A first terminal of the resistor is coupled to a signalgenerator device that is digitally programmable to generate dynamicdriving signals.

The signal generator device can be included in a digital feedback loopwhich includes means for monitoring the phase of the tank impedance andto provide corrective action, (a frequency change) for adjusting thatphase. Thus, the signal generator device can be digitally programmed toemulate the operating frequency of any previous (or future) analogproximity system and can also be digitally reprogrammed, in real time,for driving the sensing coil of the probe at one or more frequenciescorrective of any anomalous phase shift calculated from the probe ortank impedance or due to any other anomalies within the system. Forexample, the instant invention can drive the sensing coil at a precisefrequency corrective of temperature variations in the probe includingthe integral sensing coil and probe cable, and in the target.

A filter is interposed between the signal generator device and the firstterminal of the resistor to purify the output dynamic signals of thesignal generator device by eliminating, inter alia, harmonics that arecreated in the device. In addition, the filter helps reduce the noisebandwidth of the system which improves a signal to noise ratio. Thefiltered signal is driven through the resistor, extension cable (ifemployed), probe cable and coil for inducing eddy currents within thetarget. In turn, the eddy currents in the target induce a voltage in thesensing coil of the probe and hence, a change in an impedance of theprobe and extension cable (if employed) which varies as a function of,inter alia, the displacement motion and position of the target relativeto the probe.

The first and second terminals of the resistor are coupled to inputs ofa first and a second analog to digital converter respectively. In turn,the outputs of the analog to digital converters are coupled to a digitalsignal processor including a convolution means. The first analog todigital converter receives and samples: a first voltage between theserially coupled resistor, extension cable (if employed), probe cableand coil and outputs a first digital voltage signal to the digitalsignal processor. The second analog to digital converter receives andsamples the voltage between ground and the combination of the extensioncable (if employed), the probe cable and the coil and then, outputs asecond digitized voltage signal to the digital signal processor. Atiming control means is operatively coupled to the analog to digitalconverters and to the signal generator device such that the sampling issynchronously performed with the driving signal of the signal generator.This ensures, inter alia, that when the voltages are calculated therewill be exactly one cycle worth of data stored in each data set.

The digital signal processor convolves the two digitized voltages byconvolving each digitized voltage with a digital sine and cosine wave toobtain a first and a second complex voltage number. Once the convolutionof the digitized voltages is performed the impedance value of theextension cable (if employed), probe cable and coil can be calculateddirectly from the measured voltages.

The system includes an open/short/load calibration method which cancompensate for cable length included in the second electrical component.For example, the extension cable can be compensated for by using theopen/short/load calibration method according to the instant invention.Thus, the system can apply the open/short/load calibration method to themeasured impedance to obtain a compensated impedance. Furthermore, theopen/short/load calibration method can be utilized to calibrate eachprinted wire assembly within the system.

The measured impedance or the compensated impedance is then correlatedby the system to a gap value by using equations, numerical methods,algorithmic functions or lookup tables wherein gap values are correlatedto measured or compensated impedance values defining the gap or spacinginterposed between the probe and the target being monitored. This methodof measuring gap can be continuously repeated for monitoring, forexample the vibration of a rotating shaft of a machine or an outer raceof a rolling element bearing.

Additionally, the system can combine the measured impedance or thecompensated impedance value with a mathematical model value or anempirically predetermined value of an existing (or future) analogconditioning circuit that is compatible with the particular probe beingemployed. This value can be called up from a memory means associatedwith the digital signal processor. The digital signal processor combinesthis value with the measured impedance or the compensated impedance toobtain a resultant impedance defined as the tank impedance. This tankimpedance can be employed to determine the gap between the probe and thetarget by using equations, numerical methods, algorithmic functions orlookup tables wherein gap values are correlated to tank impedancevalues. Thus, the existing proximity probe can be retained and thismethod of measuring gap can be continuously repeated for monitoring, forexample the vibration of a rotating shaft of a machine or an outer raceof a rolling element bearing that was heretofore monitored by an analogeddy current proximity system.

Gap values can be outputted to a digital to analog converter forproviding analog outputs or downloaded to a processing stage for furtherprocessing and/or providing digital and/or analog outputs.

The impedance value of analog conditioning circuitry determined from themathematical model or empirically is typically dependent on operatingfrequency. Thus, once the tank impedance is determined it can be used todetermine if the system is running at the proper frequency. If thesystem is not running at the proper frequency the digital feedback loopcan be used to feedback a signal from the digital signal processor toprogram the signal generator device for dynamically adjusting thedriving signal.

Moreover, the instant invention includes a unique materialidentification method for automatically identifying a target materialand automatically calibrating itself to monitor the identified materialthereby eliminating the need for component changes and laboriousre-calibration methods inherent with prior art systems. The instantinvention also expands the unique material identification method toinclude a material insensitive method which is capable of outputting agap value substantially correct for any target material being monitoredthereby providing a material insensitive digital proximity system. Thus,the instant invention provides a digital proximity system that does notrequire component changes when being used to replace an existing systemand/or does not require re-calibration when being used with a new ordifferent target material. As a result, the instant invention provides adigital proximity system which can not be mis-calibrated when put intooperation and which eliminates the interchangeability problem found inprior art systems.

Additionally, the instant invention includes a unique inductive ratiomethod which allows a gap versus inductive ratio curve to be determinedfor a specific target material without knowing the far gap impedance ofthe probe coil and thus, without removing the probe from a machine beingmonitored. The gap versus inductive ratio curve determined by thismethod can be used to determine the gap between the probe and the targetbeing monitored. Furthermore, this method can be used to discernmoisture ingress within a probe while it is still in the machine.

OBJECTS OF THE INVENTION

Accordingly, a primary object of the instant invention is to provide anew, novel and useful method for measuring a gap between a proximityprobe and a conductive target material.

Viewed from a first vantage point, it is an object of the instantinvention to provide a method for measuring a gap between a proximityprobe and a conductive target material, the steps including: providing adatabase having a defined series of gap locus each representative of thesame gap for different target materials stored therein; measuring animpedance of a proximity probe located proximate a conductive targetmaterial to be monitored; normalizing the measured probe impedance;comparing the normalized probe impedance with the defined series of gaplocus stored in the database for determining a gap locus which defines agap between the proximity probe located proximate the conductive targetmaterial such that the defined gap is substantially correct for anyconductive target material located proximate the proximity probe forproviding a material insensitive method for measuring gaps between theproximity probe and different conductive target materials.

Viewed from a second vantage point, it is an object of the instantinvention to provide a method for measuring a gap between a proximityprobe and a conductive target material, the steps including: providing arepresentation of a defined series of gap locus each representative ofthe same gap for different target materials; measuring an impedance of aproximity probe located proximate a conductive target material, theproximity probe including a probe cable; compensating an impedancecontribution of the probe cable from the measured probe impedance todefine a measured coil impedance; normalizing the measured coilimpedance; determining a gap value between the proximity probe and theconductive target material from the normalized coil impedance and therepresentation of the defined series of gap locus wherein the gap valueis substantially correct for any conductive target material adjacent theproximity probe thereby providing a material insensitive method formeasuring gap values between the proximity probe and differentconductive target materials.

These and other objects and advantages will be made manifest whenconsidering the following detailed specification when taken inconjunction with the appended drawing figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of the digital proximity system according tothe instant invention shown employing a proximity probe including asensing coil located proximate a conductive target object to bemonitored.

FIG. 2 is a block diagram of the digital proximity system according tothe instant invention providing further detail.

FIG. 3 is a block diagram of the digital proximity system according tothe instant invention shown employing an extension cable.

FIG. 4 is a flow chart of a voltage ratio method according to theinstant invention.

FIG. 5 is a diagrammatical view of a convolution block of the digitalproximity system according to the instant invention.

FIG. 6 is a schematic of an example of analog conditioning circuitry ofan analog eddy current proximity system.

FIG. 7 is a schematic showing an equivalent circuit of the analogconditioning circuitry shown in FIG. 6.

FIG. 8 is a partial schematic of that which is shown in FIG. 7 forshowing where a feedback voltage is applied within the analogconditioning circuitry.

FIG. 9 is a general flow chart of a resonant method according to theinstant invention.

FIG. 10 is a flow chart of a cable compensation and gap measurementmethod according to the instant invention.

FIG. 11 is a block diagram of an open/short/load compensation modelincluding a four-terminal circuit block according to the instantinvention.

FIG. 12 is a diagram of a cable(s) replacing the four-terminal circuitblock shown in FIG. 11 and with the cable(s) in an open conditionaccording to the instant invention.

FIG. 13 is a diagram of the cable(s) replacing the four-terminal circuitblock shown in FIG. 11 and with the cable(s) in a short conditionaccording to the instant invention.

FIG. 14 is a diagram of the cable(s) replacing the four-terminal circuitblock shown in FIG. 11 and with the cable(s) coupled to a load accordingto the instant invention.

FIG. 15 is an exemplary graph showing a normalized impedance diagram.

FIG. 16 is a flow chart of the material identification and calibrationmethod according to the instant invention.

FIG. 17 is an exemplary graph showing normalized impedances of differentmaterials, which is employed in the material identification andcalibration method according to the instant invention.

FIG. 18 is an exemplary graph showing normalized impedances of differentmaterials and showing a series of gap locus employed by the materialinsensitive method according to the instant invention.

FIG. 19 is a flow chart of the material insensitive method according tothe instant invention.

FIG. 20 is a graph showing a normalized impedance plane of resistanceand reactance for diagrammatically defining nomenclature of an inductiveratio method according to the instant invention.

FIG. 21 is a graph showing an inductive ratio as a function of gap fordefining nomenclature of the inductive ratio method according to theinstant invention.

FIG. 22 is a flow chart of an inductive ratio method according to theinstant invention.

DESCRIPTION OF PREFERRED EMBODIMENTS

Considering the drawings, wherein like reference numerals denote likeparts throughout the various drawing figures, reference numeral 10 isdirected to the digital proximity system according to the instantinvention.

In its essence, and referring to FIGS. 1 through 4, the system 10includes, inter alia, a unique voltage ratio apparatus and method (VRmethod) for digitally measuring an unknown electrical value of anelectrical component. In one preferred form, the system 10 samples anddigitizes a dynamic voltage V₁ impressed across a serial coupling of afirst electrical component having a known electrical value and a secondelectrical component having an unknown electrical value. Additionally,the system 10 samples and digitizes a dynamic voltage V₂ impressed onlyacross the second electrical component. These two digital voltages arethen digitally convolved by the system 10 into complex voltage numbersV_(1C) and V_(2C) respectively. A ratio of the second complex number toa difference between the first and the second complex number is thendetermined by the system 10. The system 10 can use this voltage ratio todetermine the unknown electrical value of the second electricalcomponent. For example, and according to the instant invention, thesystem 10 can employ a proximity probe as the second electricalcomponent and continuously digitally measure an impedance value of theprobe monitoring, for example, a rotating shaft of a machine or an outerrace of a rolling element bearing. The digitally measured impedancevalues can then be correlated by the system 10 to displacement motionand position of the rotating shaft or the outer race of the rollingelement bearing relative to the probe for monitoring machinery status.

Specifically, and referring to FIGS. 1 through 4, the system 10 employsa proximity probe 12 which is disposed proximate a conductive ormetallic target object T (e.g., a rotating shaft of a machine or anouter race of a rolling element bearing) to be monitored. The system 10samples and digitizes a voltage V₁ impressed across a serial coupling ofa resistance means 40 having a known electrical resistance value R andthe proximity probe 12 (and, when employed, an extension cable 30coupled to the probe 12) having an unknown electrical value Z_(unknown)into a digital voltage V_(1D). Additionally, the system 10 samples anddigitizes a voltage V₂ impressed only across the proximity probe 12 (andcoupled extension cable when employed) into a digital voltage V_(2D).The system 10 then digitally convolves the two digital voltages V_(1D),V_(2D) into first and second complex voltage numbers V_(1C) and V_(2C)respectively. Then, the system 10 determines a ratio of the firstcomplex number to a difference between the first and the second complexnumber and multiplies this voltage ratio by the known electricalresistance value R to determine the unknown impedance value Z_(unknown)of the probe 12. This process follows the equationZ_(unknown)=[V_(2C)/(V_(1C)−V_(2C))]*R. The determined impedanceZ_(unknown) can then be compensated by using an open/short oropen/short/load calibration or compensation method according to theinstant invention which will be described in detail infra. The system 10then correlates the determined impedance or the compensated determinedimpedance to a gap interposed between the probe disposed proximate themetallic target object T being monitored. Iteratively repeating thevoltage ratio method results in continuously determining the unknownimpedance values of the probe which can be correlated into valuesrepresentative of the displacement motion and position of the metallictarget object T relative to the probe. Thus, the system 10 according tothe instant invention can be employed as, inter alia, a digitalproximity system for continuously monitoring rotating and reciprocatingmachinery.

More particularly, and referring to FIGS. 1 through 4, the digitalproximity system 10 includes the unique voltage ratio method (VR method)for digitally measuring the unknown electrical impedance of the probe 12operatively coupled to the system 10 and strategically coupled to amachine for sensing raw dynamic data that is correlative to the spacingbetween the probe and the conductive or metallic target object T (e.g.,a rotating shaft of a machine or an outer race of a rolling elementbearing) being monitored. The digital proximity system 10 includes theresistance means 40 having a value R, a filter means 50, a buffer, gainand offset means 60 and a signal generator means 70. The digitalproximity system 10 further includes a timing control means 80, asampling means 90, a convolution means 100 and a digital signalprocessor (DSP) means 110.

The resistance means 40 includes a first terminal 41 and a secondterminal 42 respectively coupled between a first node 44 and a secondnode 46. The proximity probe 12 includes an unknown dynamic probeimpedance having a value Z_(unknown) and is coupled between the secondterminal 42 of the resistance means 40 at node 46 and a ground node 48.Thus, the resistance means 40 and the probe 12 form a serial connection.

The probe 12 includes an integral sensing element or coil 14 and amulti-conductor probe cable 20. The sensing element 14 includes a firstelectrical lead 16 and a second electrical lead 18. The probe cable 20includes a first conductor 22 and a second conductor 24 extending from afirst end 26 to a second end 28 of the probe cable 20. The firstconductor 22 and the second conductor 24 at the first end 26 of thecable 20 are operatively coupled to the first electrical lead 16 and thesecond electrical lead 18 of the sensing element 14 respectively. Thefirst conductor 22 at the second end 28 of the cable 20 is coupled tothe second terminal 42 of the resistance means 40 at node 46 and thesecond conductor 24 is coupled to the ground node 48 thereby groundingone lead of the unknown dynamic probe impedance Z_(unknown). It isimportant to note that the configuration of the resistance means and theunknown probe impedance Z_(unknown) is neither arbitrary norinconsequential. The configuration of the instant invention grounds aconductor of the probe so as to not have both conductors varying therebyeliminating, inter alia, signal changes within the probe due to, forexample, external influences such as one moving or grabbing the cable.Furthermore, the unknown probe impedance Z_(unknown) is digitallymeasured by direct read circuitry which takes direct voltage readingsrather then inferring the voltage across or the current through theprobe.

Additionally, the instant invention can employ a multi-conductorextension cable 30 (please see FIG. 3) for extending the distancebetween the proximity probe 12 and the system 10. The extension cable 30includes a first conductor 32 and a second conductor 34 extendingbetween a first end 36 and a second end 38 of the extension cable 30.The extension cable 30 is coupled between the probe cable 20 and thesystem 10. Particularly, the first end 36 of the extension cable 30 iscoupled to the second end 28 of the probe cable 20 via a cable coupling27 such that the conductor 22 is connected to conductor 32 and conductor24 is connected to conductor 34. The first conductor 32 at the secondend 38 of the cable 30 is coupled to the second terminal 42 of theresistance means 40 at node 46 and the second conductor 34 is coupled tothe ground node 48. Thus, the extension cable 30 is coupled in serieswith the probe cable 20.

The signal generator means 70 is operatively coupled to the firstterminal 41 of the resistance means 40 at node 44 for driving a signalthrough the resistance means 40, and the probe 12 thereby impressing thefirst voltage V₁ across the serially connected resistance means 40 andprobe 12, and impressing the second voltage V₂ only across the probe 12(and extension cable if employed). Preferably, the signal generatormeans 70 is operatively coupled to the resistance means 40 at node 44via the filter means 50 and to the digital signal processor for drivinga programmable dynamic signal of one or more frequencies through thefilter means 50 and the serial connection of the resistance means/probecombination.

In particular, the signal generator means 70 preferable includes adirect digital synthesis (DDS) device 72 operatively coupled to thefirst terminal 41 of the resistance means 40 via the filter means 50 andthe buffer, gain and offset means 60 for driving the dynamic signal orwaveform through the resistance means 40 and the probe 12 (and extensioncable if employed). This dynamic signal causes the first voltage V₁ tobe impressed across the serial connection of the resistance means 40 andprobe 12 and causes the second voltage V₂ to be impressed only acrossthe probe 12 (and extension cable if employed). Typically, the sensingelement 14 of the probe 12 is strategically coupled proximate the targetto be monitored such that this dynamic signal causes the sensing element14 of the probe 12 to generate an alternating magnetic field thatinduces eddy currents in the metallic target object T. In turn, the eddycurrents in the target induce a voltage in the sensing element 14 of theprobe 12 and hence, a change in an impedance of the probe which variesas a function of, inter alia, variations of spacing or gap between theprobe and the target being monitored.

The signal generator means 70 can include a plurality of DDS devices 72coupled to the first terminal 41 of the resistance means 40 via thefilter means 50 and the buffer, gain and offset means 60 for driving aplurality of dynamic signals at different frequencies through theresistance means 40, the probe 12 and extension cable (if employed) andsubsequently performing processing including convolution as delineatedin detail infra for obtaining simultaneous impedance measurements of theprobe 12 at different frequencies which are correlative to the gapinterposed between the probe 12 and the target being monitored.

Each direct digital synthesis device 72 is preferably coupled to the DSPmeans 110 via interface 114 and generates a purefrequency/phase-programmable dynamic signal such as a sinusoidal wave.The DSP means 110 preferably includes an algorithm to program both thefrequency and the phase of the output signals which in turn can be usedto drive the probe with a frequency/phase-programmable dynamic analogsignal having an output frequency/phase which can be preciselymanipulated under full digital control. Thus, each DDS device can bedigitally programmed to output sine waves at any frequency/phase withprecision for use as driving signals or reference signals. One exampleof the DDS devices 72 is that which is manufactured by Analog Devicesand sold under part number AD9850.

The filter means 50 is interposed between the DDS devices 72 and theresistance means 40 for filtering the analog dynamic signals output fromDDS devices 72. The filter means 50 preferably includes at least one lowpass filter 52 interposed between each direct digital synthesis device72 and the first terminal 41 of the resistance means 40 to purify theoutput dynamic signals or waveforms of each synthesis device 72 foreliminating, inter alia, the harmonics that are created in the synthesisdevices 72. For example, as a result of the outputs of the directdigital synthesis devices 72 being ten plus bit digital to analogconverters, the quantitization noise needs to be filtered out using alow pass filter. Thus, the filters 52 remove the steps and smoothes outthe analog dynamic signal outputs from the DDS devices 72. Additionally,the filters 52 helps reduce the noise bandwidth of the system 10 whichimproves a signal to noise ratio. Furthermore, a half bit of noise canbe summmed in at node 54 to change the quantitization noise using adithering process. Preferably the low pass filters 52 are in the form offive pole elliptical filter devices.

Buffer, gain and offset means 60 can be interposed between filter means50 and the resistance means 40 for buffering and amplifying the analogdynamic signals and providing any desired offset of same.

The sampling means 90 is coupled to the first node 44 for sampling anddigitizing the first voltage V₁ impressed across the serially connectedresistor/probe combination. Additionally, the sampling means 90 iscoupled to the second node 46 for sampling and digitizing the secondvoltage V₂ impressed only across the probe 12 (and extension cable ifemployed). Preferably, the sampling means 90 includes a pair of analogto digital converters 92, 94 coupled to the first node 44 and the secondnode 46 respectively for sampling and digitizing the first dynamicvoltage V₁ and the second dynamic voltage V₂. The analog to digital(A-D) converters 92, 94 are preferably 14 bit, wide bandwidth convertersmanufactured by, for example, Analog Devices under part number AD9240.

The timing control means 80 provides the synchronization between theoutput signal of the signal generator means 70 and the sampling rate ofthe sampling means 90 so that the phase relationship between the outputsignal and samples is maintained. The timing control means 80 isoperatively coupled to each DDS device 72, the pair of analog to digitalconverters 92, 94, and to the DSP means 110. Thus, the DDS devices 72are clocked by the timing control means 80 so that the frequency of theoutput of these devices is very accurately set. Additionally, the timingcontrol means 80 provides the synchronization between the output of theDDS devices 72 and the sampling rate of the analog to digital converters92, 94 so that the phase relationship between the dynamic drivingsignal(s) and the sampled signals is maintained. Thus, the sampling isperformed in synchrony with the dynamic driving signals. Note that thequartz clock oscillator 84 is operatively coupled to each DDS device 72for providing a clock signal thereto.

Specifically, the timing control means 80 is an agile-clock generatorwhich preferably includes a DDS device 82 operatively coupled to andclocked by a quartz clock oscillator 84. An output of the DDS device 82is preferable filtered by a five pole elliptical filter 86 and feedbackto the DDS device 82 which then outputs a triggering signal to theanalog to digital converters 92, 94 and a timing signal to the DSP means110. The DSP means 110 is operatively coupled to the DDS devices 72, 82and may employ the timing signal sent to the DSP means 110 whendigitally programming the DDS devices 72, 82 to orchestrate thesynchronicity between the sampling rate and the dynamic signals outputby the DDS devices 72 to the probe 12. This assures that when thevoltages V_(1D) and V_(2D) are calculated there will be exactly onecycle worth of data stored per data set. Thus, the DDS devices can beused for generating the dynamic signals which excite the sensing element14 and for generating timing signals for triggering the sampling by thepair of analog to digital converters 92, 94.

The convolution means 100 can be a stand-alone device in the form of,for example, a digital down counter (DDC), that just does convolution.In this embodiment, the convolution means 100 is interposed between andcoupled to the sampling means 90 and the DSP means 110 to do theconvolution operation. The analog to digital converted values (thedigitized voltage signals V_(1D), V_(2D) which represent the dynamicvoltages V₁ and V₂ at respective nodes 44 and 46) are received andconvolved by the convolution means 100 and then supplied to the DSPmeans 110 as complex voltage numbers V_(1C), V_(2C). The advantage ofthis embodiment is that it is no longer necessary to vary the samplerate to remain synchronized to the signal generator. The DDC has thecapability of being programmed for what frequency to process. Oneexample of a commercially available digital down counter (DDC) ismanufactured by Harris Semiconductor under part number HSP 50016.

Alternatively, the digital convolution means 100 can be integrallyformed with the digital signal processor means 110 wherein the DSP means110 is operatively coupled to the pair of analog to digital converters92, 94 for receiving the first and second digitized voltage signalsV_(1D), V_(2D) from the converters and convolving the digitized voltagesinto respective complex voltage numbers V_(1C), V_(2C) via the integralconvolution means 100. Examples of DSP means 110 having integralconvolution means 100 can be found in the 210XX series of devicesmanufactured by Analog Devices. Preferably, the DSP means 110 is a40-megahertz floating point device or faster.

The process of convolving the digitized voltages into respective complexvoltage numbers V_(1C), V_(2C) via the convolution means 100 is definedas inphase and quadrature detection or quadrature synthesis.

More specifically, and referring to FIG. 5, the digitized voltagesignals V_(1D), V_(2D) can be represented by the function V cos(wT+phi)and the circles with the “X” in the middle represent digital multipliers102,104 integrally formed in the convolution means 100. The digitizedvoltage signals V_(1D), V_(2D) are each multiplied by a digital cosineand sine waveform (cos(wT) and sin(wT)) which can be pulled from a tablein memory 101, memory 112 or from memory means 120. The results of thosemultiples are accumulated and averaged or filtered by, for example, theconvolution means 100 or the digital processor means 110 to get DCcomponents. These filtered (averaged) values or transformed valuesrepresent the magnitude of real and imaginary components of the complexvoltages V_(1C), V_(2C) of the convolved digitized voltage signalsV_(1D), V_(2D). In other words, these DC components are the inphase andquadrature components of the digitized voltage signals V_(1D) and V_(2D)and represent magnitudes of real and imaginary components therebyforming complex voltage numbers V_(1C) and V_(2C) from the dynamicvoltage signals V₁ and V₂.

An alternative way of looking at this is that when you multiply a signal(e.g. either of the data strings coming out of the analog to digitalconverters tied to node 44 (V₁) or node 46 (V₂)) by a sine or a cosinewave of the same frequency, you get a DC term and a 2X AC term.Averaging the output of the multiplication then filters out the AC term.When this multiplication is performed using both a cosine and a sinefunction, you get two DC terms that represent the inphase and quadraturecomponents of the signal. These are the real and the imaginary valuesfor voltage. Note, a scaling term is usually needed after the averagingto get back to other proper engineering units. However, the term cancelsout since voltage appears in both the numerator and denominator as aresult of the instant invention using the voltage ratio method definedhereinabove.

Another possible way to implement the convolution method as contemplatedby the instant invention is to interpose a Field Programmable Gate Array(FPGA) between the analog to digital converters and the DSP means 110.The difference between this configuration and that described hereinabovefor the convolution device is that the structure of the hardwarenecessary to perform convolution method is built and programmed into theField Programmable Gate Array (FPGA).

At this point it is important to note that the idea of using digitalconvolution means 100 is paramount because if the multipliers where tobe of an analog design, the accuracy of the multipliers becomes acritical error source. For example, if one needs to distinguish a valueof 0.005 in something of magnitude 300+j100 then a required stability at1 MHz would be on the order of 1 part in 63,000. This is one to twoorders of magnitude better than you can get out of an analog multiplier.Thus, the instant invention solves this problem by using analog todigital converters for sampling the voltage signals V₁ and V₂ and thenperforming multiplication in digital format to handle the stabilityproblem. Additionally, the instant invention employs high speed analogto digital converters to avoid the additional error source (multipliergain drift).

Once the complex voltage numbers V_(1C) and V_(2C) are determined, thedigital signal processor means 110 processes the complex voltage numbersinto the unknown impedance of the probe by preferably using the voltageratio equation: Z_(unknown)=[V_(2C)/(V_(1C)−V_(2C))]*R. The DSP means100 can continuously accumulate, process and store data from theconvolution means 100 and output signals as a function of the calculatedimpedance which are correlative to the gap between the probe and thetarget (e.g., a rotating shaft of a machine or an outer race of arolling element bearing) being monitored. Particularly, the calculatedimpedance can be converted by the processor 110 into a voltage or gapvalue correlative to the spacing or gap between the probe and targetbeing monitored by using equation(s), algorithms, numerical methods orlookup tables stored in, for example, the memory means 120. It isimportant to note that the voltage ratio alone can be used to determinevalues representative of gap by accounting for the known resistancevalue within the equation(s), algorithms, numerical methods or look uptables.

Moreover, the digital signal processor can apply digital signals to thesignal generator means 70 for digitally reprogramming, in real time, thegenerator means 70 for driving the probe 12 at one or more frequenciescorrective of anomalies due to, for example: temperature variations;changes in the conductivity and permeability in the target, proximityprobe (including the integral sensing coil and probe cable) andextension cable, and anomalies due to phase shift ascertained from themeasured impedance values.

In addition to the unique voltage ratio method, the digital proximitysystem 10 includes a unique resonant method which emulates the operationof analog eddy current proximity systems thereby providing bothbackwards and forwards compatibility with existing and future analogsystems. Thus, the digital proximity system 10 provides a direct one forone replacement of existing and future analog eddy current proximitysystems.

For background, and as delineated in the background of the invention, atypical analog eddy current proximity system includes a proximity probelocated proximate a metallic target object (e.g., a rotating shaft of amachine or an outer race of a rolling element bearing) being monitored,an extension cable (if employed) and analog conditioning circuitry whichincludes a resonate circuit. The target, probe, extension cable andconditioning circuitry are all designed to interact in such a way that avoltage output from the circuitry is directly proportional to a distancefrom the probe to the target and this distance is commonly referred toas “gap”.

In general, and referring to FIG. 9, the resonant method includes thesteps of measuring the impedance of the probe and the extension cable(if employed), mirroring the impedance of analog proximity circuitry,computing a combination of the mirrored impedance with the measuredprobe and extension cable impedance (if employed), and determining a gapvalue as a function of the computed impedance which is correlative tothe spacing or gap interposed between the probe and a metallic targetbeing monitored. It should be understood that if the extension cable isnot employed the system 10 would then only measure the impedance of theprobe and compute the combination of the mirrored impedance with that ofthe measured probe impedance.

In particular, and referring to FIGS. 1 through 4, the digital Proximitysystem 10 uses the unique voltage ratio apparatus and method asdelineated in detail hereinabove to first determine a complex numberthat represents the complex impedance of any probe and extension cable(if employed) that is new or that was previously coupled to an analogconditioning circuitry input. The complex number that represents thecomplex impedance of the probe and extension cable can be held in amemory such as the memory of the DSP means 110 and/or the memory means120 which may or may not be integral with the DSP means 110.

Next, the system 10 mimics the loading that an analog conditioningcircuit puts on a tank impedance of an included resonate circuit. Thesystem 10 does this by putting a mathematical inductor and/or capacitor(mathematical Z) in parallel with the probe and extension cableimpedance (please see FIG. 3). Therefore, the system 10 accuratelymimics what happens in an actual analog conditioning circuitry foraccomplishing the important task of providing backwards compatibilitywith the existing, field installed, transducers.

One way of explaining how the system 10 mirrors or mimics the analogconditioning circuitry is by example. The schematic shown in FIG. 6shows one example of an analog conditioning circuit 170 having aresonate circuit and those having ordinary skill in the art and informedby the present disclosure should recognize the following:

First, that there is considerable circuitry in: the analog conditioningcircuitry that is in parallel with the impedance of the probe and/orextension cable at connector J2 which affects the magnitude of theimpedance and the frequency of operation.

Second, that the current driven into the tank impedance at node N_(Zp)is supplied from a collector of Q1.

Third, that the voltage at the collector of Q1 will be the tankimpedance times the current supplied or in other words, the voltage atthe collector of Q1 is directly proportional to the magnitude of thetank impedance that is to be measured. Thus, the actual value for thetank impedance (Zp) will be determined at the collector of Q1.

Fourth, that there is a transfer function from the collector of Q1 to atypical amplitude detector 172 in analog eddy current proximity systems.

Fifth, that there is a transfer function from the collector of Q1 to thefeedback of the oscillator and that this feedback affects the frequencyof operation.

Therefore, according to the instant invention, one method for mirroringor accurately mimicking analog circuitry in the digital proximity system10 includes determining an equivalent impedance of the analogconditioning circuitry, for example the equivalent impedance of theanalog conditioning circuitry 170.

The equivalent impedance may be determined by, for example, determiningthe equivalent circuit of the analog conditioning circuitry,pre-computing the impedance at different frequencies and using a look uptable (stored in memory 112 or memory means 120) to grab the appropriateimpedance value or any other method of determining circuit impedancewhich is known in the art. The first method provides the convenience ofallowing one to verify a mathematical model versus what the systemreally does while the second method is more computationally efficient.Note that empirical testing may be required to match the actual systemresponse to the mathematical model.

For example, FIG. 7 shows a therein equivalent circuit 174 of that whichis shown in FIG. 6. Some of the components therein have negligiblecontribution to the overall impedance, but it is preferred to add themfor completeness. Some other impedances that may have a minor impact oncomputing Zp are, for example, the base impedance of the transistors.

Therefore, according to the instant invention, the combination of theimpedance of the probe and extension cable (if employed) arid theequivalent impedance of the analog conditioning circuitry 170 is thencomputed. This computed value is the tank impedance (Zp) and iscorrelated by the system 10 to a gap value by using equation(s),numerical methods, algorithmic functions or lookup tables whereinimpedance values are correlated to gap values defining the gap orspacing interposed between the probe and the target being monitored.This method of measuring gap can be continuously repeated formonitoring, for example the vibration of a rotating shaft of a machineor an outer race of a rolling element bearing.

Notwithstanding, one design question that should be answered is whetherthe analog system is at the right frequency. The voltage developed atthe collector of Q1 is assumed to be directly proportional to the tankimpedance so it is exactly in phase with the voltage developed. Thisvoltage is feedback to the base of Q2 to make the analog systemoscillate, so it goes through the feedback network 176 shown in FIG. 8.

The phase shift from the node marked N_(Zp) to the feedback voltage nodeN_(Fv) is the phase delay of the oscillator. This phase shift is addedto the phase of the tank impedance and should equal the phase of theoscillator under steady-state operation. If there is an inequality, theoscillator is not at steady state and will slew towards the frequencythat satisfies that relationship. To account for this the digitalproximity system 10 computes a phase error which is defined by thefollowing equation: Phase error=phase the oscillator runs at (usually 0to 6 degrees)−Phase of the tank impedance+the phase delay in thefeedback network. Preferably, the digital proximity system 10 multipliesthis calculated phase error by a pre-computed gain term to compute howfar to adjust the frequency to mimic steady state operation of theanalog circuitry. This calculation can be computed in the DSP means 110.Thus, a digital feedback loop including the signal generator means 70can receive digital feedback signals from the digital signal processormeans 110 which are correlative to any anomalous phase error ascertainedfrom the measured impedance for digitally reprogramming, in real time,the generator means 70 for driving the sensing coil 14 of the probe 12which adjusts the frequency to mimic steady state operation of theanalog circuitry.

Thus, the digital proximity system 10 may have to account for thefrequency shift before it can compute the gap of the system.

Typically, the components used in the analog conditioning circuitry arecalled out on bill of materials (BOMs) and installed into the printedwiring assemblies (PWAs). The Zp versus gap is set when the resistorsare tweaked during calibration of the analog conditioning circuitry. Inthe digital Proximity system 10, these values are mathematicalconstructs stored in a file in the memory means 120 and pulled out asneeded to work with whatever device the system 10 happens to be pluggedinto at the time.

More specifically, and referring to FIG. 10, the resonant methodincludes measuring the unknown impedance or an uncompensated impedanceof the probe located proximate the target T and an extension cable 30(if employed) as delineated in detail supra. Next, compensation factorsor coefficients are determined from open/short or open/short/loadcalibration tables 125 stored in memory 112 or memory means 120 formedfrom an open/short or open/short/load calibration or compensation methodwhich will be described in detail infra. The measured impedance is thencompensated by using the coefficients determined from the open/short oropen/short/load calibration tables 125. The equivalent or load impedancevalue that the analog proximity circuitry would have had in parallelwith the probe and extension cable is mathematically combined with thecompensated impedance to form the “tank impedance” of the system 10. Asnoted hereinabove, the equivalent impedance values can be determined by,for example, using lookup tables 123 of empirically determined values orof mathematically modeled values. The calculated tank impedance is thenused to determine the gap the system is at. Alternatively, both thecurrent frequency and the calculated tank impedance can be used todetermine the gap the system is at by using one or more mathematicalequations 121, one or more look up tables 123, numerical methods 122 orvia algorithmic functions 124. Next, the calculated tank impedance canbe used to determine the phase shift that would have be needed to befeedback to the oscillator of the analog proximity circuitry so that itheads towards its final steady-state frequency setting. This phase shiftcan be used to adjust the frequency of the dynamic signal of the system10 which is driving the probe 12. These steps or a subset of these stepsare iteratively repeated to continuously monitor the gap between theprobe and the target being monitored.

For example, and referring to FIG. 10, the resonant method can includethe steps of measuring the uncompensated impedance of the probe locatedproximate the target T and an extension cable 30 (if employed) asdelineated in detail supra. Then, determining compensation coefficientsfrom the open/short/load calibration tables 125 stored in memory 112 ormemory means 120, compensating the measured impedance by using thedetermined coefficients and determining a gap value as a function of thecompensated impedance.

In another example, the unknown impedance of the probe and/or theextension cable can be measured at one fixed frequency to allow for thesystem 10 to be simplified. The following steps outline one methodaccording to the instant invention for measuring an unknown impedanceZ_(unknown) when using a single fixed frequency. The steps include:measuring the uncompensated impedance of the probe and extension cable(if employed); compensating the measured impedance using cablecoefficients appropriate for the single fixed frequency; determining agap value as a function of the compensated measured impedance, anditeratively repeating the measuring, compensating and determining stepsto substantially continuously measure the gap between the probe and thetarget being monitored for providing data correlative to machine status.

The coefficients appropriate for the measured impedance at the fixedsingle frequency can be empirically predetermined using, for example,the open/short or open/short/load calibration method and then stored inmemory 112 or memory means 120 and then later recalled for use with themeasured impedance value for determining the gap value as a function ofthe compensated measured impedance of the probe and/or cable.Furthermore, the equivalent or load impedance that the analog proximitycircuitry would have had in parallel with the probe and extension cablecan be combined with the compensated impedance and used in determiningthe gap value as a function of the combined impedance.

Note that each time the gap is measured using, inter alia, the resonantmethod it can be output in analog or digital form.

The cable calibration or compensation methods referred to supra will nowbe explored.

In the environment of machine monitoring the unknown impedance measuredby the digital proximity system 10 can be that of the extension cableand the proximity probe including the integral sensing element 14 andprobe cable 20. The instant invention includes an open/shortcompensation method and/or an open/short/load compensation method thatcan be employed for eliminating cable residuals (residual cableimpedance and stray cable admittance) of either the extension or probecable, or both. These two methods and the differences therebetween willbe described with the assistance of FIGS. 11 through 14 and then adetailed delineation will be presented on how these methods can beemployed for eliminating cable residuals in order to obtain true orcompensated probe impedance (Z_(probe)) and true or compensated sensingcoil impedance (Z_(sensing element)). These measured impedances orcompensated impedances can then be correlated by the system 10 to a gapvalue by using equations 121, numerical methods 122, algorithmicfunctions 124 or lookup tables 123 wherein gap values are correlated tomeasured or compensated impedance values defining the gap or spacinginterposed between the probe and the target being monitored. This methodof measuring gap can be continuously repeated for monitoring, forexample the vibration of a rotating shaft of a machine or an outer raceof a rolling element bearing.

In general, and referring to FIGS. 11 through 13, the open/shortcalibration method models the residuals (e.g., residual impedance andstray admittance) as a linear two port or four terminal networkrepresented by ABCD parameters. It should be noted that the open/shortcalibration method assumes that the network is a symmetrical network.From this restriction, the open/short compensation does not require aload measurement to know each value of the ABCD parameters. Referring toFIG. 11, a theoretical explanation and procedure are as follows:$\begin{bmatrix}V_{1} \\I_{1}\end{bmatrix} = {{\begin{bmatrix}A & B \\C & D\end{bmatrix}\begin{bmatrix}V_{2} \\I_{2}\end{bmatrix}} = \begin{bmatrix}{{AV}_{2} + {BI}_{2}} \\{{CV}_{2} + {DI}_{2}}\end{bmatrix}}$therefore;V ₁ =AV ₂ +BI ₂I ₁ =CV ₂ +DI ₂thus, the measured impedance Z is represented as: $\begin{matrix}{Z = {\frac{V_{1}}{I_{1}} = {\frac{{AV}_{2} + {BI}_{2}}{{CV}_{2} + {DI}_{2}}.}}} & \left( {{equation}\quad 1} \right)\end{matrix}$

Open measurement: when the unknown terminals 47, 49 are opened, I₂=0.Then, from equation 1, the measured impedance Z_(OM) is:Z _(OM) =A/C  (equation 2).

Short measurement: when the unknown terminals 47, 49 are shorted, V₂=0.Then, from equation 1, measured impedance Z_(S) is:Z _(S) =B/D  (equation 3).

A limited condition for the ABCD parameters is as follows: when theunknown network is “symmetric” A=D (equation 4). Thus, a symmetricalnetwork can be defined as one where the parameters A and D are equal toone another.

DUT (Device Under Test) measurement: when the DUT is connected toterminals 47, 49, its impedance value is represented as Z_(X)=V₂/I₂(equation 5).

From equations 1 and 5, its measured impedance value Z_(XM) is:$\begin{matrix}{Z_{XM} = {\frac{V_{1}}{I_{1}} = {\frac{{AV}_{2} + {BI}_{2}}{{CV}_{2} + {DI}_{2}} = {\frac{\frac{{AV}_{2}}{I_{2}} + B}{\frac{{CV}_{2}}{I_{2}} + D} = {\frac{{A\quad Z_{X}} + B}{{CZ}_{X} + D}.}}}}} & {{equation}\quad 6}\end{matrix}$

Equation 6 can be solved for Z_(x) and from equations 2 and 3, A and Bare erased to give: $\begin{matrix}{Z_{X} = {\frac{D}{C}{\frac{Z_{S} - Z_{XM}}{Z_{XM} - Z_{OM}}.}}} & {{equation}\quad 7}\end{matrix}$

From equations 2 and 4, unknown parameters C and D in equation 7 can beerased and then the true or compensated value of the unknown impedanceof the DUT can be defined as Z_(X) which is determined by an open/shortequation which is as follows: $\begin{matrix}{Z_{X} = {Z_{OM}\frac{Z_{S} - Z_{XM}}{Z_{XM} - Z_{OM}}}} & {{equation}\quad 8}\end{matrix}$wherein:

-   -   Z_(OM) is the measured open impedance    -   Z_(S) is the measured short impedance    -   Z_(XM) is the measured value of the unknown impedance        Z_(unknown) of the DUT        Note: all values are complex numbers.

In general, the system 10 can employ the open/short method as follows.First, the impedance of the probe cable 20, the extension cable 30 orboth coupled together is measured with one end left opened (FIG. 12) fordefining Z_(OM). Second, the impedance of the probe cable 20, theextension cable 30 or both coupled together is measured by shorting theone end (FIG. 13) for defining Z_(S). Third, an unknown impedance iscoupled to the probe cable 20, the extension cable 30 or both and itsmeasured impedance Z_(XM) is obtained by preferable using the voltageratio apparatus and method of the system 10 as delineated supra whereinZ_(XM) is the Z_(unknown) described hereinabove. Finally, the true orcompensated value Z_(X) of the measured unknown impedance is determinedfrom the open/short equationZ_(X)=Z_(OM)*(Z_(S)−Z_(XM))/(Z_(XM)−Z_(OM)).

For example, the true or compensated value of the impedance of theproximity probe 12 coupled to the system 10 via the extension cable 30can be calculated from the following equation:Z _(X) =Z _(OM)*(Z _(S) −Z _(XM))/(Z _(XM) −Z _(OM))wherein:

-   -   Z_(OM) and Z_(S) are respectively, the measured open and short        impedances utilizing only the extension cable 30,    -   Z_(XM) is the measured impedance of the proximity probe 12        coupled to the extension cable 30 which in turn is coupled to        the system 10 (Note that in this example Z_(XM) is the        Z_(unknown) described supra with the extension cable employed),        and    -   Z_(X) is the true or compensated impedance value of the        proximity probe 12 wherein the residuals of the extension cable        30 are eliminated.

Thus, the extension cable residuals are mathematically eliminated byusing the compensation coefficients Z_(OM), Z_(S) and Z_(XM) to definethe compensated proximity probe impedance of the proximity probe 12.

Likewise, Z_(OM) and Z_(S) can be respectively, the measured open andshort impedances utilizing only the probe cable 20 and Z_(XM) can be themeasured impedance of the sensing element 14 coupled to the probe cable20 which in turn is coupled to the system 10. Thus, Z_(X) would then bethe true or compensated value of the sensing element 14 with theresiduals of probe cable 20 eliminated.

Additionally, Z_(OM) and Z_(S) can be respectively, the measured openand short impedances utilizing both the probe cable 20 and the extensioncable 30 coupled together and Z_(XM) can be the measured impedance ofthe sensing element 14 coupled to the probe cable 20 which in turn iscoupled to the system 10 via the extension cable 30. Thus, Z_(X) wouldthen be the true or compensated value of the sensing element 14 with theresiduals of both the probe cable 20 and the extension cable 30eliminated.

Thus, the residuals of the probe cable 20, the extension cable 30 orboth coupled together can be mathematically eliminated by using thecompensation coefficients Z_(OM), Z_(S) and Z_(XM) to define thecompensated proximity probe impedance of the proximity probe 12 or thecompensated sensing element impedance of the sensing element 14.

Furthermore, true impedance values or compensation coefficients fordifferent extension or probe cable configurations including differentlengths can be stored in memory, for example, memory means 120 as lookup or calibration tables 125 and recalled when necessary. For example,the system 10 can measure the impedance of the serial coupling of theextension cable 30 and proximity probe 12 and then eliminate viaimpedance values or compensation coefficients the extension cableimpedance to determine the proximity probe impedance or eliminate viaimpedance values or compensation coefficients both the extension cableimpedance and probe cable impedance to determine the sensing element orcoil impedance. Likewise, the system 10 can measure the impedance of theproximity probe 12 and then eliminate via impedance values orcompensation coefficients the probe cable impedance to determine thesensing element or coil impedance.

The following three methods can be used to determine the values forZ_(om) and Z_(s). First, a tuned cable length is used where the lengthis set so that they most closely match known compensation values storedin the memory means of the system 10. Second, a user installs theextension cable to an input of the system 10 and then performs theopen/short compensation method as described above and then stores themeasured values in the memory means 120. Third, by mathematicallydetermining the cable length and compensation values and using thesevalues to eliminate cable residuals.

The DDS devices of the system 10 can be used for generating the drivesignal by loading digital control signals from the DSP into the DDS andletting the DDS drive the probe cable 20, the extension cable 30 or bothwith arbitrary waveforms at whatever frequency is needed.

The open/short/load compensation is an advanced compensation techniquethat is applicable to complicated residual circuits and is the preferredcable calibration or compensation method according to the instantinvention. To perform open/short/load compensation, three measurementsare required before measuring a DUT (e.g., the unknown impedance of theprobe monitoring a metallic target object). These measurements include ameasurement with the unknown terminals 47, 49 open, a measurement withthe unknown terminals 47, 49 shorted and a measurement with a standardDUT (load) having a known value coupled between the unknown terminals47, 49.

The open/short/load compensation method is particularly useful when theextension cable 30 is used whose length cannot be compensated with acable length correction function nor be minimized with the open/shortcorrection method described above.

The open/short/load correction requires the measurement data of at leastone standard DUT having a known value in addition to the open/shortmeasurement data. Similarly to the open/short method, and referring toFIG. 11, the open/short/load compensation method models the residuals asa four-terminal network circuit represented by the ABCD parameters. Eachparameter is known if three conditions are known and if thefour-terminal network is a linear circuit. $\begin{matrix}{{{From}\quad{{FIG}.\quad 11}},{Z_{1} = {V_{1}/I_{1}}},} \\{Z_{2} = {{V_{2}/I_{2}}\quad{and}}} \\{Z_{1} = {\frac{{AV}_{2} + {BI}_{2}}{{CV}_{2} + {DI}_{2}} = \frac{{A\quad Z_{2}} + B}{{CZ}_{2} + D}}}\end{matrix}$The parameters A, B, C and D can be removed when using the followingdefinitions:

-   -   Z_(O) is the measured open impedance with the terminals 47, 49        open,    -   Z_(S) is the measured short impedance with the terminals 47, 49        closed,    -   Z_(SM) is the measured impedance of the standard DUT when        connected to the terminals 47, 49,    -   Z_(std) is the true (or expected) value of the standard DUT,    -   Z_(XM) is the measured value of the DUT having the unknown        impedance and connected to the terminals 47, 49 (for example,        the proximity probe 12)    -   Z_(DUT) is the true or compensated value of the DUT having the        unknown impedance (for example, the compensated impedance of the        proximity probe 12).

The true or compensated value of the unknown impedance Z_(DUT) isdetermined by an open/short/load equation which is as follows:$Z_{DUT} = \frac{{Z_{std}\left( {Z_{O} - Z_{SM}} \right)}\left( {Z_{XM} - Z_{S}} \right)}{\left( {Z_{SM} - Z_{S}} \right)\left( {Z_{O} - Z_{XM}} \right)}$Note: all values are complex numbers.

Thus, in general, the system 10 can employ the open/short/load method asfollows. First, an impedance of the probe cable 20, the extension cable30 or both cables coupled together is measured with one end left opened(FIG. 12) and is defined as Z_(O). Second, an impedance of the probecable 20, the extension cable 30 or both cables coupled together ismeasured by shorting the one end (FIG. 13) and is defined as Z_(S).Third, Z_(SM) is defined as impedance which is measured with a knownload (a standard DUT or load) coupled to one end of the probe cable 20,to one end of the extension cable 30 or to one end of coupled cables(FIG. 14). Fourth, an impedance is coupled to the probe cable 20, to theextension cable 30 or to the combination of coupled cables and itsimpedance is measured using the voltage ratio apparatus and method ofthe system 10 as delineated hereinabove for defining Z_(XM). Finally,the true or compensated impedance of Z_(XM) is then calculated from theopen/short/load equation delineated hereinabove.

For example, the system 10 can employ the open/short/load method asfollows. First, an impedance of the extension cable 30 is measured withone end left opened (FIG. 12) for defining Z_(O). Second, an impedanceof the extension cable is measured by shorting the one end (FIG. 13) fordefining Z_(S). Third, an impedance of the extension cable is measuredwith one end coupled to a known load (a standard DUT or load) as shownin FIG. 14 for defining Z_(SM). Fourth, the system 10 is coupled to theextension cable 30 which in turn is coupled to the proximity probe 12for measuring Z_(XM). Note that Z_(XM) is the same measurement as thatof the measurement of the unknown impedance Z_(unknown) delineated suprawith the extension cable employed. Finally, the true proximity probeimpedance (Z_(probe)) can then be calculated from the open/short/loadequation as follows:$Z_{probe} = \frac{{Z_{std}\left( {Z_{O} - Z_{SM}} \right)}\left( {Z_{XM} - Z_{S}} \right)}{\left( {Z_{SM} - Z_{S}} \right)\left( {Z_{O} - Z_{XM}} \right)}$where:

-   -   Z_(O) is the measured open impedance of the extension cable 30        (FIG. 12),    -   Z_(S) is the measured short impedance of the extension cable 30        (FIG. 13),    -   Z_(SM) is the measured impedance of the load (standard load or        standard DUT) coupled to the extension cable 30 in place of the        proximity probe 12 (FIG. 14),    -   Z_(std) is the known value of the load impedance (standard load        or standard DUT),    -   Z_(XM) is the measured impedance of the probe 12 coupled to the        extension cable 30,    -   Z_(probe) is the true or compensated complex electrical        impedance of the proximity probe 12 with the residuals of the        extension cable eliminated.

Likewise, Z_(O) and Z_(S) can be respectively, the measured open andshort impedances utilizing only the probe cable 20. The impedance of theprobe cable can then be measured with a standard load coupled to one endto define Z_(SM). Next, an impedance of the probe cable can be measuredwith the sensing element or coil 14 coupled to one end to define Z_(XM).Note that Z_(XM) is the same measurement as that of the measurement ofthe unknown impedance Z_(unknown) of the proximity probe 12 delineatedsupra without the extension cable 30 employed. Finally, the true orcompensated sensing element impedance (Z_(sensing element)) can then becalculated from the open/short/load equation with the residuals of probecable 20 eliminated.

Additionally, Z_(O) and Z_(S) can be respectively, the measured open andshort impedances utilizing both the probe cable 20 and the extensioncable 30 coupled to one another and then, the impedance of the coupledprobe cable 20 and extension cable 30 is measured with a standard loadcoupled to the end of the probe cable 20 to define Z_(SM). Next, theimpedance Z_(XM) can be determined by measuring the impedance of theextension cable 30 coupled to the probe cable 20 which in turn iscoupled to the sensing element or coil 14. Thus Z_(XM) is themeasurement of the unknown impedance Z_(unknown) as delineated suprawith the extension cable 30 employed. Finally, the true or compensatedsensing element impedance (Z_(sensing element)) can then be calculatedfrom the open/short/load equation with residuals of both the probe cable20 and the extension cable 30 eliminated.

One specific method of canceling out the effects of an added extensioncable during the digital impedance measurement by the system 10 usingthe open/short/load compensation method is as follows:

-   -   1. A user cuts an extension cable 30 to the length that is most        beneficial for the mechanical installation of the proximity        probe 12 and installs a connector on the trimmed end.    -   2. The user connects the trimmed extension cable 30 to the        system 10 at nodes 46, 48 with the probe end open.    -   3. The user performs an open/short/load calibration at terminals        47, 49 as described above on the trimmed extension cable 30 and        the data is either manually or automatically stored in, for        example, memory 120.    -   4. The user connects the proximity probe 12, for example, a 0.5        or 1.0 meter proximity probe 12 to the trimmed extension cable        via connectors.    -   5. The system 10 than measures the impedance and mathematically        eliminates the residuals of the trimmed extension cable as        described supra. The residual impedance is that of the probe 12.        Thus, the system 10 is calibrated to operate with the proximity        probe 12 and outputs a linearized gap signal.

In another embodiment the system can employ a load/load/load method asfollows. First, an impedance Z_(L1) of the extension cable is measuredwith one end coupled to a first load. Second, an impedance Z_(L2) of theextension cable is measured with one end coupled to a second load.Third, an impedance Z_(L3) of the extension cable is measured with oneend coupled to a third load. Fourth, an impedance Z_(XM) is measured bythe system 10 wherein the extension cable is coupled between the digitalproximity system 10 and the proximity probe 12 and thus the systemmeasures the impedance of both the proximity probe 12 and the extensioncable 30. Finally, the true or compensated probe impedance (Z_(probe))can then be calculated from the following equation:$Z_{probe} = \frac{{Z_{std}\left( {Z_{L1} - Z_{L3}} \right)}\left( {Z_{XM} - Z_{L2}} \right)}{\left( {Z_{L3} - Z_{L2}} \right)\left( {Z_{L1} - Z_{XM}} \right)}$wherein:

-   -   Z_(L1) is the measured impedance of the extension cable 30        coupled to a first load,    -   Z_(L2) is the measured impedance of the extension cable 30        coupled to a second load,    -   Z_(L3) is the measured impedance of the extension cable 30        coupled to a third load (standard load or standard DUT),    -   Z_(std) is the known value of the third load impedance (standard        load or standard DUT),    -   Z_(XM) is the measured impedance of the coupled probe 12 and        extension cable 30,    -   Z_(probe) is the compensated complex electrical impedance of the        proximity probe 12.

Thus, the proximity probe impedance of the proximity probe is calculatedas a function of the measured impedance, the first load impedance, thesecond load impedance, and the third load impedance for compensating forextension cable residuals. The proximity probe impedance is thencorrelated to a gap between the proximity probe and the conductivetarget material.

Preferably, the second load has an impedance that is less than theimpedance of the first load and the third load has an impedance that isless than the impedance of the first load and greater than the impedanceof the second load.

Additionally, note that this method can also be used to determine thetrue value of the sensing element wherein Z_(L1), Z_(L2), Z_(L3)respectively replace Z_(O), Z_(S) and Z_(SM).

It should also be noted that the following conditions may be requiredfor the open/short and/or the open/short/load methods. First, whengetting the open correction data, the distance between measurementterminals should be the same as the distance that is required foractually holding the DUT. Secondly, when getting the short correctiondata, the measurement terminals should be shorted or connected to ashorting device and the residual impedance should be less than theimpedance value of the DUT. Thirdly, when selecting a standard DUT(load) for the load correction step there is no restriction that aninductor must be used for inductance measurement, or capacitor must beused for a capacitance measurement. Any device can be used if itsimpedance value is accurately known. It is important to use a stablestandard DUT not susceptible to influences of environment such astemperature or magnetic fields. From this viewpoint, capacitors orresistors are better suited than inductors which are more susceptible tothe environment. When measuring a DUT's various impedance values, a 100to 1,000 ohm resistor standard DUT (load) may provide the best results.When measuring a DUT of one impedance value, a standard DUT (load)having approximately the same impedance as that of the DUT to bemeasured may provide the best results. A 100 to 1,000 ohm resistorstandard DUT (load) may provide the best results for a DUT having anunknown impedance which is either very high or very low.

A standard DUT (load) may be measured using a direct connection to nodes46, 48 of the system 10 after performing the open/short correctionmethod at nodes 46, 48 to determine compensation coefficients to be usedin calculating the value of the standard load using equation eight (8)as delineated hereinabove. Additionally, It should be noted that theopen/short/load compensation method may be required to employ a stableknown load which is measured in the same way that the DUT will bemeasured and is of the same approximate value.

In conclusion, the system 10 can be calibrated at nodes 46, 48 by usingthe open/short/load method thereby defining a first calibration plane atnodes 46, 48 and including in memory 120 a correction or compensationfunction or table of the internal impedance of the system. For example,the calibration tables 125 can include a correction function orcompensation including compensation coefficients which compensate forthe resistance means 40 such that only the ratio V_(2C)/(V_(1C)−V_(2C))of the complex voltages is required to determine the unknown impedancevalue Z_(unknown) of the probe 12. Additionally, when an extension cableis coupled to nodes 46, 48 the system 10 can be calibrated at the end 36of the extension cable 30 by using the open/short/load method therebydefining a second calibration plane. Thus, the calibration tables 125can include a correction or compensation function which compensates forthe extension cable such that the ratio V_(2C)/(V_(1C)−V_(2C)) of thecomplex voltages is correlative to unknown impedance value Z_(unknown)of the probe 12 and not the extension cable 30 and probe 12 combination.

Additionally, the system 10 can be used to make all of the abovedelineated impedance measurements for the open/short method, theopen/short/load method and the load/load/load method and store thesevalues in memory.

Referring to FIGS. 16 through 22, the digital proximity system 10further includes a unique material identification and calibrationmethod, a unique material insensitivity method, and a unique inductiveratio method.

In order to understand how these unique methods work, one must firstunderstand a “Normalized Impedance Diagram”. Referring to FIG. 15, anormalized impedance diagram is shown and is comprised of a multiplicityof normalized impedance curves. This graph can be generated by taking aproximity probe and measuring its impedance at different frequencies andat different gaps from, for example, a standard E 4140 steel target. Thelines 182 through 197 radiating outward (from the (0.0, 1.0) points) aregap lines. They represent the normalized impedance due to the target ata constant frequency as the gap is changed from very close (therightmost ends of the lines) to the farthest gap (the 0.0, 1.0 point).These lines rotate clockwise along arrow F as the frequency isincreased. The arcs 200 through 208 represent the impedance of the probelocated at a fixed gap as the frequency varies.

The method of obtaining this basic normalization is as follows:

-   1. Measure a far gap impedance of the probe: Far gap    impedance=R_(o)+jwL_(o).-   2. Measure an impedance of the probe near the target: Near gap    impedance=R+jwL.-   3. Determine a normalized impedance which is comprised of a    normalized resistance term and a normalized reactance term as    follows:    -   Normalized resistance=(R−R_(o))/wL_(o) and    -   Normalized reactance=wL/wL_(o.)-   4. Plot each point on a graph and connect the points done at the    same frequency.-   5. Connect the points done at the same gap thereby obtaining a graph    as shown in FIG. 10.

Each target has its own characteristic normalized impedance diagram andit has been observed that the curves rotate clockwise as theconductivity and permeability of the target increase. Also, it has beenobserved that there is much more reactive change with gap than there isa resistive change as the conductivity and permeability of the targetincrease. The prior art systems are much more sensitive to resistancechanges than to reactive changes which makes the prior art systems moredifficult to calibrate for high conductivity materials as a result ofthere not being much resistance change to detect.

Note that the basic normalization method can be followed to measure thefar gap and the near gap impedance of the probe in combination with theextension cable to obtain a “Normalized Impedance Diagram” of theprobe/extension cable combination.

Additionally, one or more normalized impedance curves can be generatedby taking a probe and measuring its impedance at different frequenciesand different gaps with different target materials and storing thisinformation in, for example, the memory means 120.

Now with the basic normalization method in mind, the unique materialidentification and calibration method, the unique material insensitivitymethod and the unique inductive ratio method will be delineated indetail according to the instant invention.

Referring to FIGS. 16 and 17, the unique material identification andautomatic calibration method of the instant invention allows the system10 to identify a material that the system 10 is monitoring and toautomatically calibrate the system 10 for that material.

More specifically, and referring to FIG. 16, the material identificationand automatic calibration method of the digital proximity system 10includes the following steps. At the outset, the probe of the digitalproximity system 10 is located proximate the target material to beidentified and the system 10 measures the impedance of the probe asdelineated in detail supra. The system 10 then determines the normalizedimpedance value of the probe by using this measured impedance which maybe compensated according to the instant invention and the far gapimpedance of the probe which, for example, can be manually entered viainput means 148 or called up from memory means 120. Next, the normalizedimpedance value can be correlated to a point or impedance on apreviously stored normalized curve for a specific target material andprobe type combination. Once the correlation is found, the material ofthe target can be identified. Alternatively, a user can enter thematerial type into the system 10 via an input means 148 which would thencorrelate a stored normalized curve to the specific target material andprobe type combination.

Note a multiplicity of normalized impedance curves for a multiplicity ofdifferent targets and probe types may be stored in the memory means 120as, for example, material identification tables 126. These tables canthen be accessed at any time by the system 10 for identifying thematerial the probe is monitoring and performing the self calibrationprocess based on the identified material. These curves can be previouslygenerated by taking one or more probes and measuring their impedancevalues at different frequencies and different gaps with different targetmaterials and storing this information in, for example, the table 126 ofthe memory means 120.

Additionally, the memory means 120 can include calibration data 134which includes the parameters necessary for automatically calibratingthe system 10 for the identified material. Therefore, once the materialof the target is identified, the digital proximity system 10 can pullthe appropriate system calibration data out of the memory means 120 forautomatically calibrating the system 10 for the identified material andthus, generating gap data for monitoring the identified material.

FIG. 17 shows an example of a normalized impedance diagram 210 whichreflects normalized impedance curves for the same probe running at asingle frequency, but looking at multiple targets. Each line 212 through220 radiating outward from the (0.0, 1.0) point represents thenormalized impedance of different target materials at the constantfrequency as the gap is changed from very close (the rightmost end ofthe line) to the farthest gap (the 0.0, 1.0 point). This information canbe stored in memory 120 and the material identification and automaticcalibration method can be used to determine point 222 and correlate thepoint to a normalized curve of a 4140 type of material. This informationcan then be used to determine the calibration parameters from thecalibration data stored in the memory 120 for automatically calibratingthe system 10 and generating, or this example, a gap of 33 mils.

Note that the aforementioned material identification and automaticcalibration method can be followed with the probe being replaced withthe probe/extension cable combination to obtain a normalized impedancevalue which is correlated to a point on a previously stored normalizedcurve for a specific target material. Once the correlation is found, thematerial of the target can be identified.

This method has the huge advantage of being backwards compatible withanalog proximity systems.

As mentioned hereinabove, and referring to FIGS. 16 and 17, the uniquematerial identification method of the instant invention allows thesystem 10 to identify a material located proximate the proximity probe20.

In one form, the system 10 digitally measures the complex impedance ofthe probe 20 disposed adjacent one or more different yet known targetmaterials and driven at one or more different frequencies. The system 10then calculates a normalized impedance curve for each different targetmaterial at each different frequency and stores these curves as anequation, as an algorithmic function or as a database of values in amemory, for example, memory means 120. Thereafter, the system 10subsequently identifies unknown materials by first digitally measuringthe complex impedance of the probe 20 driven at one or more differentfrequencies and disposed adjacent an unknown material. Next, the system10 calculates the normalized impedance curve for the unknown material atthe one or more different frequencies by using an equation analgorithmic function or a database of values for the unknown material ateach driving frequency. The system 10 then compares the equation (oralgorithm) determined for the unknown material to one or more previouslydetermined equations (or algorithms) for known materials to obtain orinterpolate a match for identifying the unknown material. Alternatively,the system 10 compares one or more values in the database of values forthe unknown material at each driving frequency to one or more values inone or more previously determined databases of known materials to obtainor interpolate a match for identifying the unknown material. Note thatall measured impedances can be compensated by using the open/short oropen/short/load compensation methods according to the instant inventionprior to the probe or coil impedance being normalized.

In a second forms the system 10 identifies a unknown material bydepending primarily on the curve shape of the normalized impedanceresponse and the angle of the vector the normalized impedance responsesweeps out from the normalized resistance value 0.0 and the normalizedreactance value 1.0 to the actual normalized impedance value of thetarget at any particular frequency. Thus, material identification is notbased on the absolute position of any normalized impedance reading andas a result, lines do not have to overlap to indicate the same materialcharacteristics. Variations in “liftoff” or separation between the probeand the target may cause variations in the absolute position of thenormalized impedance measurement, but will not affect its curve orangular relationship with the origin.

When different materials have very similar conductivity the system 10can measure the impedance characteristics of each target at two or moredifferent gaps to determine the “liftoff line” for each material. Thisassists in identifying the relative position of the material'snormalized impedance response when plotted on a graph.

The system 10 can utilize these forms alone or in combination with thematerial identification and automatic calibration method for monitoringrotating and reciprocating machinery as explained hereinabove and withreference to FIGS. 16 and 17.

Another use of the material identification methods or forms according tothe instant invention is in the area of identifying and/or sorting coinsand precious metals. For example, the system 10 can determine thenormalized impedance of a series of coins and/or precious metals(targets) placed adjacent the probe 12. The normalized impedance valuescan be recorded at several different frequencies. These values can thenbe compared or plotted with known standards 123 of coins and/or preciousmetals for material identification, and discrimination.

Material discrimination is based primarily on the curve shape of thenormalized impedance response and the angle of the vector the normalizedimpedance response sweeps out from the normalized resistance value 0.0and the normalized reactance value 1.0 to the actual normalizedimpedance value of the target(s) at any particular frequency. Thus,material discrimination is not based on the absolute position of anynormalized impedance reading and as a result, normalized curves do nothave to overlap to indicate the same material characteristics.Variations in “liftoff” or distance between the probe and any target maycause variations in the absolute position of the normalized impedancemeasurement, but will not affect its curve or angular relationship withthe origin.

When different materials have very similar conductivity the system 10can measure the impedance characteristics of each target at twodifferent gaps to determine the “liftoff line” for each material. Thisassists in identifying the relative position of the material'snormalized impedance response when, for example, plotted on a graph.

One empirical example of the discrimination method delineated above usedthe system 10 to preform eddy current analysis on different metal andcoin types. The system 10 measured the impedance of the different metaland coin types, normalized these impedances and plotted the results. Itwas shown that a curve shape of the normalized impedance response and anangle of the vector or curve shape of the normalized impedance responsethat sweeps out from the normalized resistance value 0.0 and thenormalized reactance value 1.0 for gold, gold coins, silver, silvercoins and copper-nickel silver dollars was such that the discriminationof one from the other was easily discernable with a single impedancemeasurement. These materials were also easily discriminated againstplatinum and palladium.

In the case of distinguishing between platinum and palladium, differentmaterials having very similar conductivity, the system 10 measured theimpedance characteristics of each target at two different gaps todetermine the “liftoff line” for each material. This assisted inidentifying the relative position of the material's normalized impedanceresponse when, for example, plotted on a graph.

Referring to FIGS. 18 and 19, The unique material insensitive method ofthe instant invention allows the system 10 to monitor different targetmaterials without having to be re-calibrated for each different materialthereby providing a material insensitive digital proximity system 10.

More specifically, and referring to FIG. 18, a graph is shown of anormalized impedance diagram for the system 10 running at a singlefrequency, but looking at multiple targets. Each line (230 through 238)radiating outward from the (0.0, 1.0) point represents the normalizedimpedance of different target materials at a constant frequency as thegap is changed from very close (the rightmost end of the line) to thefarthest gap (the 0.0, 1.0 point). These lines rotate clockwise as theconductivity and permeability of the target increase. The arcuate lines240, 242 and 244 are a series of locus which connect the points on eachline (230 through 238) that are at the same gap. One or more normalizedimpedance curves each including a series of locus can be generated bytaking a probe and measuring its impedance at different frequencies anddifferent gaps with different target materials and storing thisinformation in, for example, a database or table(s) 128 of the memorymeans 120. Each locus can be represented by an equation(s) 129 ornumerical methods 130 which approximate the arcuate lines of constantgap. Thus, the system 10 is designed so that any impedance lying on thelocus of constant gap would output the same gap reading. No end userinteraction is necessary in this method.

In one form, and referring to FIGS. 18 and 19, the material insensitivemethod of the digital proximity system 10 includes the steps of:determining a plurality of normalized impedance curves as delineatedsupra for different materials and preferably storing the curves in adatabase of the memory means; defining a series of locus lines on theimpedance curves that represent the same gap for the differentmaterials; measuring an impedance of the probe located proximate atarget material to be monitored, normalizing the measured probeimpedance and comparing the normalized probe impedance with the seriesof locus (the arcuate lines) stored in the database for determining agap locus that corresponds to the normalized impedance value of theprobe wherein the corresponding gap locus reveals a gap valuesubstantially correct for any target material being monitored therebyproviding a material insensitive digital proximity system 10.

In another form, the material insensitive method of the digitalproximity system 10 includes the steps of: determining a plurality ofnormalized impedance curves as delineated supra for different materialsand defining a series of locus lines on the impedance curves thatrepresent the same gap for the different materials; storing anequation(s) or numerical methods which approximate the arcuate locuslines in the memory means; measuring an impedance of the probe locatedproximate a target material to be monitored, normalizing the measuredprobe impedance and using the equations(s) or numerical methods fordetermining a gap locus that corresponds to the normalized impedancevalue of the probe wherein the corresponding gap locus reveals a gapvalue substantially correct for any target material being monitoredthereby providing a material insensitive digital proximity system 10.

Note that the material insensitive method described hereinabove can befollowed when employing a probe/extension cable combination in place ofthe probe only.

Specifically, The step of measuring the impedance of the probe locatedproximate a target in the two former material insensitive methods canfurther include measuring the impedance of the probe and an extensioncable wherein all the subsequent steps are carried out using thismeasured combination of impedance in place of just the probe impedanceand all the previous steps are carried out using a probe/extension cablecombination.

In yet another form, and the material insensitive method of the digitalproximity system 10 can include the step of mathematically estimatingthe sensing element or coil impedance of the probe by removing anycontribution of impedance from the integral probe cable and theextension cable (if employed). Any contribution of impedance from theintegral probe cable and the extension cable (if employed) can bedetermined from the open/short/load calibration method delineated above.

Referring to FIGS. 20 through 22, the unique inductive ratio method ofthe instant invention allows a normalized impedance curve to bedetermined for a specific target without knowing the far gap impedanceof the probe coil and thus, without removing the probe from a machinebeing monitored.

As delineated supra, the far gap impedance is needed to actuallydetermine the normalized impedance of the probe and to develop thenormalized impedance curve for a specific target which in turn can beused to determine the gap between the probe and the target beingmonitored. There is a normalized impedance value for each target at eachfrequency and gap.

Experiments have shown that normalized impedance diagrams generated fromdifferent probes in the same series of transducers have very littlevariation. This is because the normalizing process used to generate thediagrams removes the variations caused by differences in resistance andinductance between the different coils. All that remains is the probegeometry and the target material. Coil geometry is very consistent inregards to how the target interacts with the coil. In fact, water in acoil will cause very little error in the normalized impedance diagramuntil the probe gets so wet it's almost shorted out.

Unfortunately, one can not use this technique directly to measure probegap because it depends on knowing the far gap impedance of the probecoil, which can not be determined without removing the coil from amachine being monitored by the probe. Accordingly, there is a need for amethod and apparatus which solves these problems.

Referring to FIGS. 20 through 22, the digital proximity system 10includes the unique inductive ratio method which is based on thenormalized impedance response, but is independent of the unknownvariables.

In general, and assuming that the probe is mounted in a machine at anunknown gap, an initial step of the inductive ratio method is to measurethe impedance of the probe at two different frequencies f₁ and f₂(please see FIG. 20). Thus, the impedance at f₁ is X1=(R1+jwL1) and theimpedance at f₂ is X2=(R2+jwL2). Next, the instant invention assumesthat the far gap impedance at f1 is r1+jwl1 and the far gap impedance atf2 is r2+jwl2. Then, the normalized impedance is calculated as follows:

For X1: R1n = (Rl − r1)/w1l1 w1L1n = w1L1/w1l1 For X2: R2n = (R2 −r2)/w2l2 w2L2n = w2L2/w2l2

As noted above the resistance is unreliable and therefore, the focuswill be on the reactance measurement. In addition, it was noted suprathat the far gap is unknown and as a result, w1L1 n and w2L2 n can notbe calculated because l1 and l2, inductances at far gap, are unknown.

However, applicant has discovered that a function can be defined toremove the unknown variables. Specifically, applicant has discoveredthat if a function is defined to equal a normalized reactance at f₁divided by a normalized reactance at f₂ the unknowns can be made todisappear. This function can be defined as, for example, an inductiveratio function γ. Therefore γ=(w1L1 n/w2L2 n) and from the equations forX1 and X2 hereinabove we have the following:(w 1 L 1 n/w 2 L 2 n)=(w 1 L 1/w 1 l 1)/(w 2 L 2/w 2 l 2) and thus,(w 1 L 1 n/w 2 L 2 n)=(w 1 L 1/w 2 l 2)*(w 2/w 1)*(l 2/l 1),wherein the first term on the right side of the equation can be measuredby the system 10 or an impedance meter; the second term is the knownfrequencies and the third term is the inductances at far gap whichchanges very little over frequency and therefore can be approximated asequaling one.

This function corresponds to using the change in inductance between twofrequencies at one gap to determine the actual gap. A curve 240 of thefunction γ versus gap can be precomputed and the value determined in themeasurement above can be used to estimate the gap of the probe inquestion. Graphically, this is shown in FIG. 21 wherein γ which isdepicted as γ(g) is plotted versus gap. In other words, the gap is equalto the measured reactance at f1 divided by measured reactance at f2times the frequency at f2 divided by the frequency at f1. Therefore, theinstant invention provides a method (generally depicted in FIG. 22) thatdefines a function of gap that is primarily a function of probe geometrywithout it's actual inductance or resistance being a major factor.

Moreover, the inductive ratio method can further include the step ofmathematically removing the effect of the integral cable of the probe onthe impedance of the sensing coil, mathematically removing the effect ofthe extension cable on the impedance of the probe, or mathematicallyremoving the effect of both the integral cable of the probe and theextension cable on the impedance of the sensing coil using the cablecalibration methods described hereinabove.

For example, the inductive ratio method can further include the step ofmathematically removing the effect of both the integral cable of theprobe and the extension cable on the impedance of the sensing coil(using the cable calibration methods described hereinabove) to obtainonly the impedance of the coil after initially measuring the impedanceof the probe at two different frequencies f₁ and f₂. As a result, theimpedance at f₁ defined as X1=(R1+jwL1) and the impedance at f₂ definedas X2=(R2+jwL2) hereinabove can also be used to define the impedances ofonly the sensing coil at two different frequencies. Thus, the methoddefined supra can be identically carried out with the addition of thiscable compensation step.

Specifically, the inductive ratio method can include the followingsteps. First, measuring the uncompensated impedance of the probe andextension cable (if employed) at two different frequencies f₁ and f₂.Second, determining compensation factors from open/short/loadcalibration tables. Since this method can be generally done at fixedfrequencies, the compensation factors may be pre-computed. Third,compensating the measured impedances using the coefficients determinedfrom the open/short/Load method. Fourth, mathematically removing thecable effect(s) on measured impedance so that only the sensing coilimpedance remains. Note that in a system that is trimmed for thismethod, the cables will be physically trimmed so that they match asclosely as possible the cable compensation values programmed into thesystem 10. Fifth, computing the function γ, which is the reactance ofthe coil at one frequency divided by the reactance of the same coil at adifferent frequency. Sixth, determining gap from the value γ, andseventh, iteratively repeating the previous six steps.

The advantage of the inductive ratio method according to the instantinvention is that many of the variables that affect the impedancemeasurement are eliminated using this method. Some of the variableseliminated are: the probe resistance, the probe inductance, the value ofthe known resistance in the detector, the voltage magnitude and phasedriven into the known resistor and the reference driving the analog todigital converters.

As noted, this method determines gap in a way that is very insensitiveto the series resistance of the coil. This is important because mostprobe system failures cause a change in series resistance, but not incoil inductance. Errors like: loose connectors, temperature variations,and most significantly water in the probe all cause a change inresistance, but little or no change in reactance. The only single endederror sources left are the probe geometry and the reference driving thedigital to analog output.

Additionally, this method can be used as a way of detecting the gap ofprobe that is installed in the machine, detecting the gap of a probethat may be contaminated with water thereby precluding the resistiveterm of impedance from being used and/or detecting the gap of a probewherein a far gap impedance can not be estimated to normalize themeasured impedance of the probe. Furthermore, this method can be used toprovide redundant measurements of gap in the digital proximity system 10thereby providing a cross check of actual system performance.

In operation, and referring to the drawings, the proximity probe 12 istypically coupled proximate a target to be monitored, for example, arotating shaft of a machine or an outer race of a rolling elementbearing for monitoring the gap therebetween. Therefore, the probe isstrategically coupled to the machine for sensing raw dynamic data thatis correlative to the spacing between the probe and the target of themachine being monitored to obtain a signature of the status of themachine.

The dynamic voltages V₁ and V₂ are continuously converted into periodsor cycles of digital samples. The periods or cycles of digital samplesare subsequently convolved into corresponding periods or cycles ofcomplex voltage numbers V_(1C) and V_(2C) which are used to determinedynamic impedance values of the probe 12. The dynamic impedance valuesare typically correlated to gap values correlative to the displacementmotion and position of the conductive target material being monitoredrelative to the probe. The impedance or gap values may be outputted toan analog output via a digital to analog converter 142. The analogoutput may be in the form of alarms, circuit breakers, etc. Thesedevices are set to trip when the analog output is outside a user setnominal operating range.

The impedance or gap values may be outputted to a host computer 146and/or to a processor 160 for further processing for the use ofmonitoring rotating or reciprocating machinery. In one form, andreferring to FIGS. 1 and 3, the digital proximity system 10 includes acommunications link 144, for example, a serial communications channel orinterface which is operatively coupled to the digital signal processormeans 110 for outputting signals to the host computer 146 and thus, toan end user. The serial communications channel 144 allows the digitalsamples or the convolved signals to be outputted from the digital signalprocessor means 110 to the remote computer 120 without sending the fulldynamic information of the original signals V₁ and V₂ thereby providingan important advantage of reducing the bandwidth of the communicationsignals. Additionally or alternatively, the impedance or gap values maybe outputted to the processor 160 where the values may be continuouslyaccumulated, processed and/or stored and, at any time, can betransmitted to the host computer 146 for further processing and/oroutput to an end user for the use of monitoring rotating orreciprocating machinery.

Furthermore, the digital signal processor means 110 or the processor 160may perform signal reduction on the digitized impedance or gap valuesand then output that information to the remote computer 120 via theserial communications link 118 and/or directly to the digital to analogconverter 116. For example, the digital signal processor means 110 orthe processor 150 may perform signal reduction in the from of peak topeak amplitude detection, DC gap detection, nX amplitude and phasedetection and/or spectral content detection.

Moreover, memory means 120 can include an EEPROM tied to the DSP suchthat when the DSP first powers up it loads operating information andparameters stored in EEPROM into its internal memory 112. The EEPROM canbe replaced with a dual port RAM (DPRAM) and as far as the DSP isconcerned it looks like an EEPROM. In one form, the processor means 160is coupled to the DPRAM and in turn, DPRAM is coupled to the DSP. Thus,when the DSP is held in recess the processor means 160 can loadprogramming and data into the DPRAM and once the DSP is released the DSPpulls everything out from the DPRAM and into internal memory 112.

The system 10 provides timely, meaningful and actionable information toend users. The behind the scenes activities that the system 10 mayperform to verify its own condition and validate its data is a processwhich is not one task or idea, but a process by which the system 10self-validates. The system 10 enables some level of additionalself-checking over existing systems. It is these aspects of theaforementioned process, which are as follows:

1. The system 10 can self identify target materials (or designed to workwith all metals) thereby resulting in a system 10 which be cannotmis-calibrated when put into operation.

2. Multiple signal processing algorithms may be run at the same time onthe system 10. This allows cross-checking the different methodsdescribed above to verify proper system operation with the same eddycurrent probe. As an example, the inductive ratio method can be used tohelp tell if a probe is wet while it is still in the machine.

3. The system 10 input bandwidth is sufficiently high to be able todetect intermittent connections on the probe and extension cable. Forexample, an open or short will cause a sudden change in voltage at theanalog to digital converters. This change is faster than a rotor canmove to cause a change in analog to digital readings. Thus, by checkingthe slew rate of the signal we can check to see if it is faster than therotor can move. If it is to fast, the cause must be an electrical faultlike an intermittent connector. If the bandwidth was too slow, we couldnot differentiate the problems.

4. The memory allows the system 10 to know if it has an intermittent onits input power wiring and the digital communications channel or linkallows it to communicate its problems to an asset management softwaresystem at the host. As an example, the system 10 know that it reset dueto power glitches three times in the last hour and digitallycommunicates that to an asset management system (Host computer) whichcould check to see if the power had actually been turned off. If not,there is trouble with the wiring.

5. The digital communications channel and the memory associated with theDSP and/or the CPU allows the system to generate it's own maintenancerequests.

6. The memory associated with the DSP and/or the CPU, and the digitalcommunications channel allows the system to store a complete record ofit's own checkout following installation. This allows the system to beable to communicate back up to the management software whether or notit's been subject to a loop check, when that occurred and the results ofthat test when it was run.

7. The memory associated with the DSP and/or a CPU, and the digitalcommunications channel allows the system to be used with a portablecheckout device that includes a bar code reader for recording serialnumbers on probes and extension cables. This allows the configurationsoftware to upload which probe is tied to which extension cable from thesystem 10. Standard labels could also be provided indicating bearingnumber and X,Y, spare X, spare Y to link these into the system 10. Thishelps eliminate translation errors caused by having a user write thedata down and then keypunching them into the system 10.

8. Including the complete signal chain in the system 10 software(including serial numbers) allows remote access for a product servicegroup to look for configuration errors or trace repairs that may need tobe done. It also allows spares to be ordered without having to havesomeone go and look at the installation.

9. The barcode technology can work the other way when the systems 10 isdisconnected and re-assembled. The portable device can request theprobe/extension cable serial numbers from the system 10 and then make atone or a beep representing good or bad as the technician scans barcodeslooking for the right one to tie into the system.

10. The system 10 can include a signal processing algorithm that isessentially immune to gap and very sensitive to material condition (sortof an electrical runout measurement system). This can be used to createa waveform representing the material condition of the shaft. Thispattern may be compared between X-Y pairs to help verify that the probeorientation and direction of rotation are all configured correctly.

11. The system 10 can stop driving the probe and extension cable, butstill measure the voltage developed across it due to ground loops orRFI. This can be done during system assembly as a check.

12. The system 10 can measure the wideband RMS voltage and compare thatto the one frequency that the system 10 is measuring at to see if noiseis being injected into the signal for some reason. Note, the system 10may not be getting bad readings because of the narrow bandwidth and thesystem 10 is still able to detect that the signal noise is there. Thiscan be correlated to the “bump in the night” data that may cause somekind of glitch. This is very similar to a NOT 1X measurement made,however the system can discern exactly how much synchronous signal isbeing driven through the system so any NOT 1X will be due to harmonicdistortion (which should not change unless a hardware problem occurs)and outside noise. If necessary, one could compute spectra of the signaland separate out harmonic distortion (an internal hardware problem) withinternally or externally generated noise. It's also possible to computethe spectra using different sampling frequencies and figure out theexact frequency that's causing trouble (assuming it's not widebandnoise). This is because one will be able to identify where the foldoverfrequencies occur and can identify the aliased frequencies.

13. The 1X signal may also be used to help verify probe orientation anddirection of rotation.

14. An internal timestamped event list may be maintained in the systemto document when changes were made to it's configuration. This can beused to help verify that there were no NOT OK times from the time ofsome recent event back to its last verification cycle.

Furthermore, the system 10 also provides a solution to a need for moresystems to be used as references (working standards) duringmanufacturing process of analog systems. The stability of the system 10increases at a result of at least the following three reasons: One, thesystem 10 design is inherently more stable because it depends on theratiometric measurements used in the Analog to Digital (A-D) converters,rather than on the bias through a PN junction operating on, for example,a 1 MHz signal. Two, the tank inductor in the analog system has beeneliminated and replaced with a mathematical equation. The tank inductoris the most sensitive component and has a tendency to “walk” over time.“Walk” refers to a ferrite core inductor's tendency to experience shortand long term drift in its impedance value. It is not known what causesit, but it is known that it's there. Three, it is possible to performopen/short/load calibration on every working standard system 10 at thebeginning of the day or work order to re-zero it's response.

Moreover, having thus described the invention, it should be apparentthat numerous structural modifications and adaptations may be resortedto without departing from the scope and fair meaning of the instantinvention as set forth hereinabove and as described hereinbelow by theclaims.

1. A method for measuring a gap between a proximity probe and aconductive target material, the steps including: providing a databasehaving a defined series of gap locus each representative of the same gapfor different target materials stored therein; measuring an impedance ofa proximity probe located proximate a conductive target material to bemonitored; normalizing the measured probe impedance; comparing thenormalized probe impedance with the defined series of gap locus storedin the database for determining a gap locus which defines a gap betweenthe proximity probe located proximate the conductive target materialsuch that the defined gap is substantially correct for any conductivetarget material located proximate the proximity probe for providing amaterial insensitive method for measuring gaps between the proximityprobe and different conductive target materials.
 2. The method of claim1 wherein the step of providing the database having the defined seriesof gap locus each representative of the same gap for different targetmaterials includes using a lookup table to represent the defined seriesof gap locus.
 3. The method of claim 1 wherein the step of providing thedatabase having the defined series of gap locus each representative ofthe same gap for different target materials includes using at least oneequation to represent the defined series of gap locus.
 4. A method formeasuring a gap between a proximity probe and a conductive targetmaterial, the steps including: providing a representation of a definedseries of gap locus each representative of the same gap for differenttarget materials; measuring an impedance of a proximity probe locatedproximate a conductive target material, the proximity probe including aprobe cable; compensating an impedance contribution of the probe cablefrom the measured probe impedance to define a measured coil impedance;normalizing the measured coil impedance; determining a gap value betweenthe proximity probe and the conductive target material from thenormalized coil impedance and the representation of the defined seriesof gap locus wherein the gap value is substantially correct for anyconductive target material adjacent the proximity probe therebyproviding a material insensitive method for measuring gap values betweenthe proximity probe and different conductive target materials.
 5. Themethod of claim 4 wherein the determining step further includescomparing the normalized coil impedance with the representation of thedefined series of gap locus in the database for determining a gap locusthat corresponds to the normalized impedance value of the coil fordefining the gap value between the proximity probe and the conductivetarget material wherein the gap value is substantially correct fordifferent conductive target materials adjacent the proximity probethereby providing the material insensitive method for measuring gapvalues between the proximity probe and different conductive targetmaterials.